Apparatus and method for an integrated photodiode in an infrared receiver

ABSTRACT

A method and apparatus are shown for integrating a photodiode and a receiver circuit on a single substrate. An input signal is received with the photodiode. The receiver circuit is configured to suppress feedback from an output terminal of the receiver circuit to the photodiode by amplifying the input signal to produce an amplified input signal, controlling the gain of the input signal amplification responsive to the magnitude of the amplified input signal, comparing the amplified input signal to a detection threshold voltage to produce a digital data signal, and holding the gain at a substantially constant level in response to a fast signal transition in the digital output signal.

This is a continuation-in-part of application Ser. No. 08/827,402, filedMar. 27, 1997, now U.S. Pat. No. 5,864,591.

BACKGROUND OF THE INVENTION

1. Field of Invention

The present invention relates to an integrated photodiode and infraredreceiver circuit.

2. Description of the Related Art

Infrared wireless data communication is a useful method for short range(in the approximate range of 0-10 meters) wireless transfer of databetween electronic equipment; such as, cellular phones, computers,computer peripherals (printers, modems, keyboards, cursor controldevices, etc.), electronic keys, electronic ID devices, and networkequipment. Infrared wireless communication devices typically have theadvantages of smaller size, lower cost, fewer regulatory requirements,and a well defined transmission coverage area as compared to radiofrequency wireless technology (i.e. the zone of transmission is boundedby physical walls). In addition, infrared wireless communication hasfurther advantages with regard to reliability, electromagneticcompatibility, multiplexing capability, easier mechanical design, andconvenience to the user as compared to cable based communicationtechnology. As a result, infrared data communication devices are usefulfor replacing 0-10 meter long data transfer cables between electronicdevices, provided that their size and costs can be reduced to that ofcomparable cable technology. As examples of the type of wirelesscommunications links that are presently in use, the Infrared DataAssociation (IrDA) Physical Layer Link Specification 1.1e specifies twomain physical layer infrared modulation protocols.

The IrDA Physical Layer Link Specification 1.1e also specifies two modesfor modulation of data on the infrared transmitted signal. One mode is alow-speed (2.4 Kbp/s to 115 Kbp/s) on-off infrared carrier usingasynchronous modulation where the presence of a pulse indicates a 0 bitand the absence of a pulse indicates a 1 bit. The second mode is a highspeed (576 Kbp/s to 4 Mb/s) synchronous Four Pulse Position Modulation(4 PPM) method in which the time position of a 125 nS infrared pulse ina 500 nS frame encodes two bits of information. The 1.1e specificationalso specifies a preamble pattern which is sixteen repeatedtransmissions of a predetermined set of symbols.

Infrared data communications devices typically consist of transmitterand receiver components. The infrared data transmitter section consistsof one or more infrared light emitting diodes (LEDs), an infrared lens,and an LED current driver. A conventional infrared data receivertypically consists of an infrared photodiode and a high gain receiveramplifier with various signal processing functions, such as automaticgain control (AGC), background current cancelling, filtering, anddemodulation. For one-directional data transfer, only a transmitter atthe originating end and a receiver at the answering end is required. Forbi-directional communication, a receiver and transmitter at each end isrequired. A combined transmitter and receiver is called a transceiver.

A representative example of a conventional infrared data transmitter andreceiver pair is shown in FIG. 1A. Infrared transmitter 10 includes LED16 which generates a modulated infrared pulse in response to transistor14 being driven by the data signal input at D_(IR). The modulatedinfrared signal is optically coupled to an infrared detector, such asphotodiode 24 normally operated in current mode (versus voltage mode)producing an output current which is a linear analog of the opticalinfrared signal falling on it. The infrared pulses generated by LED 16strike photodiode 24 causing it to conduct current responsive to thedata signal input at D_(IR) thereby generating a data signal received atD_(IR).

In receiver 20, the signal received at D_(IR) is transformed into avoltage signal V_(IR) and amplified by amplifier 26. The signal outputfrom amplifier 26 then feeds into comparator 42 which demodulates thereceived signal by comparing it to a detection threshold voltage V_(DET)in order to produce a digital output data signal at D_(OUT).

The received signal waveform will have edges with slope and will ofteninclude a superimposed noise signal. As a result, V_(DET) is ideallyplaced at the center of the received signal waveform so that the outputdata signal has a consistent waveform width despite the slope of thereceived signal edges. Also, placing V_(DET) at the center of thereceived signal improves the noise immunity of receiver 20 because thevoltage difference between V_(DET) and both the high and low levels ofthe received signal is maximized such that noise peaks are less likelyto result in spurious transitions in D_(OUT).

The received signal, however, can vary in amplitude by several orders ofmagnitude due primarily to variations in the distance betweentransmitter 10 and receiver 20. The strength of the received signaldecreases proportional to the square of the distance. Depending on therange and intensity of the infrared transmitter, the photodiode outputssignal current in the range of 5 nA to 5 mA plus DC and AC currentsarising from ambient infrared sources of sunlight, incandescent andfluorescent lighting. As a consequence, the center of the receivedsignal waveform will vary, whereas V_(DET) must generally be maintainedat a constant level. To address this problem, receivers typicallyinclude an automatic gain control (AGC) mechanism to adjust the gainresponsive to the received signal amplitude. The received signal is fedto AGC peak detector 36 which amplifies the signal and drives currentthrough diode 32 into capacitor 28 when the signal exceeds the AGCthreshold voltage V_(AGC) in order to generate a gain control signal.The gain control signal increases in response to increasing signalstrength and correspondingly reduces the gain of amplifier 26 so thatthe amplitude of the received signal at the output of amplifier 26remains relatively constant despite variations in received signalstrength.

At a minimum, infrared receiver 20 amplifies the photodetector signalcurrent and then level detects or demodulates the signal when it risesabove the detect threshold V_(DET) thereby producing a digital outputpulse at D_(OUT). For improved performance, the receiver may alsoperform the added functions of blocking or correcting DC and lowfrequency AC ambient (1-300 uA) signals and Automatic Gain Control (AGC)which improves both noise immunity and minimizes output pulse widthvariation with signal strength.

The structure of the conventional discrete PIN photodiode 24 isillustrated in FIG. 1B. A wafer 50 is lightly doped with N dopant inorder to produce an intrinsic region 56. A P+ region 52 is formed on onesurface of the wafer and an N+ region 58 is formed on the opposingsurface of wafer 50 with intrinsic region 56 interposed P+ region 52 andN+ region 58. A reflective layer 60, typically gold, is disposed on thesurface containing P+ region 58 with reflective layer 60 also serving asthe electrical contact to N+ region 58. A metal contact 54 is disposedon the surface containing P+ region 52 to provide the electricalconnection to the P+ region.

Typically, one power supply potential is applied to the reflective layer60 and another power supply voltage is applied to contact 54 to reversebias the PN junction formed by P+ region 52 and N+ region 18. This formsa depletion region within the intrinsic region 56 wherein electron andhole charge carrier pairs generated by light photons incident upon theintrinsic region 56 are rapidly accelerated toward the P+ and N+ regionsrespectively by the electric field of the reverse bias voltage. Chargecarrier pairs are also typically generated outside the depletion regionwithin intrinsic region 56 which diffuse, due to random thermal motionof the carriers, at a much slower velocity until they reach either thedepletion region or the junction formed by P+ region 52 and intrinsicregion 56 of photodiode 24.

A conventional photodiode that is designed for high quantum, i.e. lightconversion, efficiency requires that the light path within the photocurrent collection zone, i.e the depletion and non-depletion zoneswithin intrinsic region 56, be sufficient in length so that most of thelight photons of the incident light signal area absorbed and convertedinto electron-hole pairs that are collectable at the P+ and N+ regions.Usually, this requires that the width of the intrinsic region 56, whichis the primary light collection region, be several times the lengthrequired for light absorption. If diode 10 has an efficient back-sidereflector, such as reflective layer 60, which effectively doubles thelight path within diode 24, then the intrinsic region 56 of thephotodiode can be made narrower. For a typical near infrared siliconphotodiode, the nominal absorption path length is about 15-25 microns.The path length should be at least two to three times the nominalabsorption path length to obtain good light conversion efficiency.

The inclusion of lightly doped intrinsic region 56 between the P+ and N+regions 52 and 58 results in a PIN photodiode with a wider depletionregion, depending on the magnitude of the reverse bias voltage, whichimproves the light collection efficiency, increases speed, and reducescapacitance over that of a simple PN diode structure.

The PIN photodiode is typically produced by diffusing the N+ region 58on the back side of the lightly doped (N) wafer 50, diffusing the P+region 52 on the topside of the wafer 50, and then adding metal contactsto each side of the wafer. Typically, the backside contact areaconnected to N+ region 58 is reflective layer 60 and is made of gold.The reflective layer is then typically connected to the ground voltageterminal.

Although a PIN photodiode outperforms a standard PN diode, the PINphotodiode structure cannot be easily manufactured by standardsemiconductor processes wherein fabrication is typically performed ononly one side of the semiconductor wafer 50.

In typical high volume applications, it is now standard practice tofabricate the receiver circuitry and transmitter driver in a singleintegrated circuit (IC) to produce a transceiver IC. As described above,it is difficult to integrate an efficient PIN photodiode on the samesemiconductor substrate as the transceiver circuit. As a result, adiscrete infrared photodiode is typically assembled with the transceivercircuit and an LED, along with lenses for the photodiode and LED, into aplastic molded package to form a transceiver module. The transceivermodule is designed to be small in size and allow placement in theincorporating electronic device so as to have a wide angle of view,typically through an infrared window on the transceiver casing. Thetransceiver IC is designed to digitally interface to some type of serialdata communications device such as an Infrared Communication Controller(ICC), UART, USART, or a microprocessor performing the same function.

Accordingly, it is advantageous to integrate a photodiode and receiveror transceiver circuit on a single substrate to reduce the overall sizeof the resulting infrared device and to reduce the costs involved withassembling a discrete photodiode with a receiver or transceiver chip toproduce a receiver or transceiver module.

SUMMARY OF THE INVENTION

The present invention relates to an apparatus and method for integratinga photodiode and a receiver circuit on a single substrate.

An embodiment of an integrated photodiode and receiver circuit on asubstrate, according to the present invention, includes a first diffusedregion in the substrate for receiving an input signal, a first circuitinput terminal coupled to the first diffused region, and a circuitoutput terminal. An input amplifier is interposed between the firstcircuit input and receiver output terminals, where the input amplifierreceives and amplifies the input signal to produce an amplified inputsignal, and wherein the input amplifier varies the gain of the inputamplifier responsive to a gain control signal. A bandpass filter isinterposed between the input amplifier and the circuit output terminal,where the bandpass filter receives and bandpass filters the amplifiedinput signal so as to produce a filtered input signal. A comparator isinterposed between the bandpass filter and the circuit output terminaland compares the filtered input signal to a detection threshold voltagelevel in order to generate a digital output signal. A delay circuit isinterposed between the comparator and the circuit output terminal whichreceives the digital output signal and generates a delayed digitaloutput signal responsive thereto. An automatic gain control circuitreceives the filtered input signal and compares the filtered inputsignal to an automatic gain control threshold voltage and generates thegain control signal responsive thereto. An isolation control signalgenerator receives the delayed digital output signal and generates anisolation control signal responsive thereto. An isolation switch isinterposed between the input amplifier and the automatic gain controlcircuit, which receives the isolation control signal and, responsivethereto, isolates the automatic gain control circuit from the amplifiedinput signal.

Another embodiment of an infrared receiver circuit formed on asubstrate, according to the present invention, includes a firstdiffusion in the substrate and an input amplifier having first andsecond input terminals, a gain control terminal and an output terminal,where the first input terminal receives a first bias voltage and thesecond input terminal is coupled to the first diffusion. A bandpassfilter has input and output terminals, where the input terminal iscoupled to the output terminal of the input amplifier. A comparator hasfirst and second input terminals and an output terminal, where the firstinput terminal receives a detection threshold voltage, the second inputterminal is coupled to the output of the bandpass filter, and the outputterminal is coupled to an output terminal of the receiver circuit. Anautomatic gain control circuit has first and second input terminals andan output terminal, where the first input terminal is coupled to theoutput of the bandpass filter, the second input terminal receives anautomatic gain control threshold voltage, and the output terminal iscoupled to the gain control terminal of the input amplifier. A delaycircuit has input and output terminals, wherein the input terminal iscoupled to the output terminal of the comparator such that an outputdata signal is generated at the output terminal of the delay circuitresponsive to the input data signal. An isolation control signalgenerator has input and output terminals, wherein the input terminal iscoupled to the output terminal of the delay circuit, and wherein theisolation control signal generator generates a pulse of predeterminedduration at its output terminal responsive to an edge in the output datasignal. An isolation switch has input, output and control terminals,where the control terminal is coupled to the output terminal of theisolation control signal generator such that the isolation switchisolates the input terminal thereof from the output terminal thereofresponsive to the automatic gain control signal, the input terminal ofthe isolation switch is coupled to the output terminal of the inputamplifier, the output terminal of the isolation switch is coupled to thefirst input terminal of the automatic gain control circuit such that theisolation switch is interposed between the first input terminal of theautomatic gain control circuit and the output terminal of the inputamplifier whereby the isolation switch isolates the automatic gaincontrol circuit from the output terminal of the input amplifierresponsive to the isolation control signal.

An embodiment of a method for suppressing feedback in a photodiode and areceiver fabricated on a substrate, according to the present invention,involves receiving an input signal with the photodiode, amplifying theinput signal to produce an amplified input signal, controlling the gainof the input signal amplification responsive to the magnitude of theamplified input signal, comparing the amplified input signal to adetection threshold voltage to produce a digital data signal, andholding the gain at a substantially constant level in response to a fastsignal transition in the digital output signal.

The features and advantages of the present invention will become morereadily apparent from the following detailed description of a preferredembodiment of the invention which proceeds with reference to theaccompanying drawings.

BRIEF DESCRIPTION OF THE DRAWINGS

In the following drawings, elements which are identical or analogousbetween drawings are identified with reference numbers which are alsoidentical or analogous.

FIG. 1A is a circuit diagram of a conventional infrared transmitterreceiver pair.

FIG. 1B is a cross-sectional diagram of a conventional PIN photodiode.

FIG. 2 is a diagram of an embodiment of an integrated receiver andphotodiode according to the present invention.

FIG. 3 is a functional block diagram of one embodiment of the infraredreceiver circuit of FIG. 2.

FIG. 4 is a functional block diagram of another embodiment of theinfrared receiver circuit of FIG. 2.

FIG. 5 is a functional block diagram of yet another embodiment of theinfrared receiver circuit of FIG. 2.

FIG. 6 is a functional block diagram of still yet another embodiment ofthe infrared receiver circuit of FIG. 2.

FIG. 7 is a functional block diagram of a further embodiment of theinfrared receiver circuit of FIG. 2.

FIG. 8 is a diagram of an embodiment of an integrated receiver andphotodiode of the present invention utilizing two photodiodes and adifferential input receiver.

DETAILED DESCRIPTION OF THE PRESENT INVENTION

Because of the cost associated with assembling the transceiver moduleinto a single package, it is desirable to integrate the photodiode andthe receiver circuit on a single silicon substrate. However,constructing an IrDA receiver or transceiver with an integratedphotodiode has typically not been either cost effective or sufficientlyoptically sensitive to have the range to meet the IrDA lowspeedspecification.

In general, IrDA receivers typically use discrete photodiodes with areasof 3.4 to 25 square millimeters in order to produce the minimum signallevel, typically 30 nA to 220 nA, required by the minimum detectthreshold of an IrDA receiver at a range of 1 meter. A receiver ortransceiver IC typically has an area of 2 to 10 square millimetersdepending upon the IC fabrication process technology employed and thecomplexity of the circuit design.

Although photodiodes can be constructed using a standard IC fabricationprocess, the resulting photodiodes typically suffer from reducedefficiency, signal bandwidth, or very high capacitance as compared tophotodiodes fabricated using a process optimized for photodiodes.Because there are significantly fewer processing steps and fewerphoto-lithographic masks of low optical resolution required for aphotodiode optimized process, a photodiode specific silicon processtypically costs about one half or less per unit area than a modernsilicon IC process. In addition, because process yields are higher andtesting is simpler for a photodiode specific process, there is a furtherreduction in cost per area of perhaps another 20% to 40%. However, theprocessing and yield cost advantages of the photodiode specific processare partially offset by the assembly cost incurred in the handling,inventory, die attachment, and bonding of the discrete photodiode into areceiver or transceiver module. Also, using conventional design andfabrication techniques, the photodiodes in an IrDA receiver ortransceiver are significantly larger than the receiver or transceiver ICand the photodiodes can be fabricated as discrete devices at lower costper unit area. Thus, it is more cost effective not to integrate thephotodiodes with the receiver or transceiver, despite the savingsobtained by eliminating the photodiode assembly cost.

Furthermore, photodiodes constructed using a standard IC process willhave a light conversion efficiency per unit area which is typically 20%to 80% of the efficiency of a photodiode fabricated using a photodiodespecific process. Utilizing the assumption that the photodiodeprocessing costs are typically one half that of the IC process, theintegrated diode must be less than half the area of an externalphotodiode in order for an integrated photodiode and receiver ortransceiver IC to be more cost effective than a discrete photodiodesolution. The integrated diode will therefore have a signal output thatis 10% to 40% the signal output of a cost-equivalent discretephotodiode.

There are still other problems with photodiodes fabricated usingstandard IC processes. One major problem is that a photodiode fabricatedon a standard IC process typically has a capacitance per unit area thatis over 100 times higher than the capacitance per unit area of aphotodiode fabricated using a process optimized for photodiodes. Thisincreased capacitance causes a significant reduction in the infraredreceiver noise/bandwidth performance which in turn requires the use of alarger integrated photodiode than would otherwise be required.

The smallest photodiode signal current which can be detected, given aparticular receiver bandwidth, must be greater than the equivalent inputnoise level of the receiver. Hence, if the receiver's noise level can bereduced then a smaller received signal can be detected. The lower boundon an IC receiver's equivalent input noise level is limited by the inputtransistor noise performance and area, supply voltage, input device biascurrents, and the source impedance which is determined by signalfrequency and photodiode capacitance. The principles of low noisephotodiode amplifier design, which govern the input noise parameters,are well known to those skilled in the art and will not be expanded uponhere. However, given a particular bandwidth, it should be understoodthat as the capacitance of the photodiode increases and its impedancedecreases, then the equivalent input noise current will increase as asquare root function, assuming that the infrared receiver amplifier isoptimized for minimum noise under these conditions.

For example, a four times increase in the input capacitance of thephotodiode will typically result in a two times increase in theequivalent input noise current of a receiver including the photodiodewhen the receiver is optimized for minimum noise within the samebandwidth. Notice, however, that increasing the area of a photodiode byfour times increases the output current gain by four times although itonly increases the input noise by two times. Consequently, increasing aphotodiode area increases its effective gain, or signal-to-noise ratio,for a given signal illumination level. However, it is also clear that,for a photodiode which has a capacitance that is 100 times greater perunit area, the receiver input noise current of an optimized receiver isincreased by 10 fold and therefore requires the minimum detect thresholdof the receiver to be increased by 10 fold. Thus, although a photodiodeconstructed from a standard IC process may have only a moderatelydegraded efficiency as compared to a photodiode constructed using aphotodiode optimized process, the 100 fold larger capacitance of thestandard IC process photodiode degrades the receiver noise performanceso as to effectively reduce input sensitivity by 10 fold.

Another problem is that an efficient photodiode typically uses thesubstrate as one of the diode contacts because the light absorptiondepth of the silicon typically exceeds by several times the shallowjunctions on standard IC processes. Since the substrate contact istypically P material, this means that the substrate is an anodeconnection, which is the reverse of standard external photodiodestructure which usually have cathode connections to the substrate.Consequently, an integrated photodiode receiver typically needs to beable to function with the anode to substrate, which is usually coupledto a ground potential supply rail.

In view of these constraints, it is nonetheless possible to build anIrDA receiver with sufficient sensitivity and bandwidth to operate usinga small photodiode integrated on the same IC. As an example, such areceiver might use a cost effective integrated photodiode of about 2 mm²in area. Assuming that the integrated photodiode has about 60% of theefficiency of a standard discrete photodiode, the integrated photodiodewould output about a 18 nA pulse when suitably lensed and illuminatedwith 4 uW/cm² of infrared light, which is the minimum infraredirradiation level specified for IrDA slow speed operation with a 10⁻⁸bit error rate. The receiver utilizing the integrated photodioderequires an equivalent input noise level over the pulse bandwidth ofabout 2 nA and a signal detection threshold of about 9 nA.

A major problem for the design of IrDA infrared receivers is outputfeedback noise. The integrated photodiode is vulnerable to feedbackcapacitance from the receiver output D_(OUT) to the photodiode. Forexample, an integrated photodiode of 2 mm² will have a feedback couplingcapacitance in the range of 2 to 10 femto Farads (fF) between the uppersurface of the photodiode and the surface of the receiver output bondingwire, which will typically be less than 2 mm distance away. Although theintegrated photodiode is smaller than a discrete external photodiode,the integrated photodiode is necessarily located closer to the receiveroutput because it is on the same IC as the receiver circuit.Consequently, the feedback coupling capacitance is of the same order ofmagnitude as an external photodiode. In addition, because the receivermust be more sensitive to compensate for the smaller and less efficientintegrated photodiode, the effects of feedback are proportionatelylarger.

Specifically, taking the above examples, if the feedback capacitance isas little as 2 femto Farad and the receiver output has a 5V swing, then10 femto Coulombs (fC) of charge will be transferred to the receiverinput terminal for each transition at the receiver output terminal. Ifthe receiver bandwidth is 600 kHz, which is a typical value for a 115Kbps lowspeed IrDA receiver, then the time constant of the signaltransient is (1/600 kHz)/6.28 or about 265 nsec. Consequently, the 10 fCof charge will have an apparent current amplitude of (10 fC/265 nsec) or376 nA. Notice that this is over 40 times the minimum detect thresholdof 9 nA in the above example and will cause disruption of the inputsignal over 80% of the operating range of the receiver.

Some of the conventional remedies for output to input feedback problemsof IrDA receivers are: 1) to use a large photodiode so that the outputfeedback is proportionately small compared with the signal; 2) installshielding between the output and the input photodiode; and 3) balanceddifferential output lines and/or balanced input photodiodes. All ofthese solutions entail increased cost either due to increases in totalphotodiode area, increased package complexity, or the cost of adding ashield.

In addition, for an IrDA receiver or transceiver IC incorporating anintegrated photodiode, adding a shield between the photodiode and thereceiver output is especially difficult. Since the IC is very small,typically about 3 to 4 mm long, the shield needs to be placed accuratelywithin 1 mm after die attach and wire bonding. This extra assemblyprocess adds cost to the final package. Another difficulty is that theshield must be designed not to block the infrared light falling on theintegrated photodiode.

Another receiver output to photodiode input feedback mechanism notpresent with an external photodiode but which may be present with aninfrared receiver integrated with an on-chip photodiode is feedbackcoupling through the substrate. This may be due to either carriersinjected by the receiver circuitry or by distributed RC coupling throughthe substrate. For this feedback coupling mechanism, external shieldingwill not work, although some benefit can be had by placing diffused Pand N collector rings connected to suitable supply voltages. These acteither as limited substrate shields or as collectors for substratecarriers.

Finally, another problem for infrared data receiver performance, whetherbuilt with an external or an integrated photodiode, is external noisepickup, most notably from adjacent digital signal lines. Although themagnitude of the external noise signal is typically less than themagnitude of the feedback signal from the receiver's own outputterminal, the external noise can cause receiver disruption from digitallines or other close-by circuit nodes with significant AC signal levels.External shields are often used to control noise from external signals,which again increases the cost of the package and/or increases its totalsize.

FIG. 2 illustrates a first embodiment of a receiver module 200,according to the present invention, composed of a receiver circuit 220fabricated on the same IC substrate 210 as a photodiode 230. Thephotodiode 230 is constructed by forming an N-diffusion 232 formed inP-substrate 210. The N-diffusion 232 is connected to the input terminalD_(IR) of the receiver 220 and the P-substrate 210 is grounded.Optionally, a transparent conductive shield 240 is formed overN-diffusion 232 and the shield is also grounded. The transparentconductive shield can be formed of any transparent conductive material,such as a polysilicon conductive layer. However, a drawback of includingthe transparent conductive shield 240 is that it doubles the capacitanceto ground of photodiode 230. The output D_(OUT) of receiver 220 has afeedback capacitance 40 to photodiode 230.

FIG. 3 illustrates an embodiment of a receiver circuit 420 designed tomitigate the effects of feedback through control of the feedback phasewith the design of the receiver bandwidth filter design such thatreceiver circuit 420 is suitable for use as the receiver circuit of FIG.2. For infrared receivers which demodulate using on-off modulation (suchas the modulation specified by IrDA), it is possible to receive signalssignificantly below the feedback transient amplitude provided that thefeedback is in phase with the received signal. This is accomplished bydesigning the receiver so that the feedback from the data output ispositive such that the feedback actually reinforces the received signalbecause the polarity of the feedback spike corresponds to the polarityof the received signal. If the receiver transient response has littleovershoot, and either no AGC or high signal threshold AGC is used, thenthe positive feedback acts as dynamic hysteresis, producing an outputpulse without any spurious transitions.

The infrared receiver 420 of FIG. 3 is designed so that the feedbackthrough parasitic capacitor 440 from D_(OUT) to the photodiode inputD_(IR) is positive. Thus, a negative transition in the received signalat D_(IR) results in a negative transition in the data output signal atD_(OUT) which will, in turn, generate a corresponding negative spike inthe feedback signal.

The received signal is countered by the much larger amplitude of thefeedback spike which causes the signal at the input of detect comparator442 to cross the detect threshold V_(DET) repeatedly. This results inmultiple transitions in data output signal thereby corrupting the data.

When the receiver is designed for positive feedback, as is receiver 420,the positive feedback reinforces the received signal. The positivefeedback, when combined with the received signal at D_(IR), results in asignal at the input of comparator 442 that swings farther away fromV_(DET) responsive to the edges in D_(IR) generating a single outputpulse at D_(OUT).

Bandpass filter 434 must be designed for good damped transient responseto suppress signal ringing and overshoot. An example of a suitablefilter is a Gaussian bandpass filter, which has rolled-off edges in thesignal output from the filter that have low ringing and overshoottransient response. In addition, the filter will temporally spread outthe energy contained in the feedback spike. For example, a 20 nsec.spike is transformed into a 300 nsec. spike, which further contributesto the dynamic hysteresis discussed above.

However, in order to prevent AGC desensitization, the AGC threshold VAGCof receiver 420 needs to be set above the peak feedback value by amargin adequate to prevent the peak feedback value from causing the gainto be adjusted downward by the AGC. There are undesirable consequencesof a high AGC threshold compared with a low AGC threshold. First, AGCnoise quieting (which reduces signal interference from noise) is lesseffective. Secondly, the output pulse width will vary more with signallevel.

AGC reduces the front end gain of amplifier 426 responsive to anincreasing input signal on D_(IR). Generally, in the absence of an inputsignal, amplifier 426 will be highly sensitive because the AGC permitsthe gain to be high. In the presence of noise, however, the AGC willreduce the gain in response to the input signal including the noiselevel. This improves the noise immunity of the receiver 420. Thereceiver will function so long as the received signal strength isgreater than the amplitude of the noise. From a practical standpoint, ina noisy environment, this permits the sending and receiving devices tobe moved closer together to strengthen the received signal and thecommunications link will be able to function. In the absence of AGC, thenoise level will prevent the receiver from capturing the transmittedsignal without corruption of the output data signal even when thetransmitting device and receiver are close together because the noisesignal will still have high enough amplitude to cause spurious outputtransitions in between valid output transitions.

Also, AGC improves the fidelity of the pulses in the data outputD_(OUT). Despite careful filter design, some ringing, overshoot andundershoot will still occur in the receiver. AGC reduces the effect ofringing, overshoot and undershoot when it reduces the sensitivity ofamplifier 426. Further, because there is also ramping on the receivedwaveform which can cause widening or narrowing of the signal pulseunless the detect threshold V_(DET) is in the center of the waveform,AGC improves the fidelity of the pulse by maintaining V_(DET) at thecenter of the waveform.

The amplitude of the positive feedback, however, also causes the AGC toadjust the receiver sensitivity downward. As the AGC reduces the gain inresponse to the positive feedback, the sensitivity of the receiver tothe transmitted signal is also reduced and, particularly at high signalpulse rates, can cause the receiver to lose the input signal.

Whereas the receiver circuit 420 of FIG. 3 is effective in reducing theeffects of feedback, it still suffers from the effect of feedbacktransient overshoot or ringing, which, if it exceeds the detect levelV_(DET), will cause undesirable extraneous output pulse transitions.Although the use of well known filter design techniques can limittransient overshoot to a negligible level, in practice, reducing it to avalue below ⅕ or {fraction (1/10)} the peak level is difficult due tovariable phase shift effects both within and outside the infraredreceiver. Some of these variable phase shift effects are due to normalvariances in parameters such as transmit pulse shape, photodiode timeconstant, photodiode capacitance, receiver output load capacitance,receiver supply voltage, and filter component values.

Receiver 420 can beneficially decrease the disruptive effects offeedback by 10 db-20 db for infrared receivers used with edge triggered,serial data communication controllers which do not need an accurate datapulse width or with receiver systems which do not require the benefitsof a low threshold AGC.

Another embodiment of a receiver suitable for use in the presentinvention is shown in FIG. 4. Infrared receiver 620, which is suitablefor the use of low threshold AGC, adds delay 650 to the output signalfrom comparator 642, typically delaying the output by ½ of the pulseinterval of the data signal to permit decay of the feedback pulse.Receiver 620 also includes a signal disable switch 648 controlled by AGCdisable one-shot 652 which blocks the signal to the AGC input upon theleading edge of an output pulse transition on D_(OUT).

By delaying the output signal, the peaks of the feedback signal fromD_(OUT) to D_(IR) are shifted in time so that the feedback peak occurstoward the center of the pulse in the received signal at D_(IR). Also,AGC disable one-shot 652 generates a signal disable pulse responsive tothe falling edge in the output signal at D_(OUT) which causes AGCdisable switch 648 to open and isolate the input of AGC peak detector636 from the received signal path for the duration of the signal disablepulse. The signal disable pulse must persist for a time intervalsufficient for the feedback transient to settle below levels which wouldcause AGC gain reduction. AGC peak detector 636 is therefore isolatedfrom the signal path at the time that the feedback pulse appears at thenegative input to comparator 642. As a result, the gain control voltagestored in capacitor 628 during the disable period reflects the receivedsignal strength and is not corrupted by the feedback signal.

Alternatively, the AGC disable one-shot 652 may be replaced with anoutput edge triggered disable one shot which will generate a disablepulse responsive to both the falling and rising edges of the outputsignal at D_(OUT). An edge triggered disable will isolate the AGC peakdetector 636 during feedback pulses for both the edges of the outputpulse at D_(OUT). This approach has the advantage that larger levels offeedback can be tolerated or the use of a low AGC threshold voltagelevel is permissible because the gain control voltage is not adverselyaffected by the feedback pulse from the rising edge of the output signalat D_(OUT).

Whereas the performance of receiver 620 is substantially better than theperformance of conventional receivers, it requires that receiver 620 bedesigned so that the feedback from D_(OUT) to D_(IR) is positive.Negative feedback will still cause spurious transitions in the outputsignal because the feedback is coupled to the input of comparator 642.

Receiver 820 of FIG. 5 shows yet another embodiment of a single inputreceiver circuit suitable for use as the receiver 220 in the presentinvention but where the design of receiver 820 is not dependent uponpositive feedback from output terminal D_(OUT) to input terminal D_(IR).Signal disable switch 848 is positioned between the output of bandpassfilter 834 and the inputs of both comparator 842 and AGC peak detector836. Output edge triggered disable one-shot 852 receives the delayedoutput signal at D_(OUT) and generates a disable pulse responsive toeach of the falling and rising edges of the output signal. The disablesignal must persist for a period long enough for the feedback transientsto decay below a level that would cause spurious transitions in theoutput signal from comparator 842. The disable pulses cause signaldisable switch 848 to isolate comparator 842 and AGC peak detector 836from the received signal path during the times when the falling andrising feedback peaks are present at the output of bandpass filter 834.This configuration permits receiver 820 to obtain the improved AGCperformance of receiver 620. However, receiver 820 is not dependent uponpositive feedback because the input of comparator 842 is isolated fromthe feedback peaks, thereby preventing the negative feedback peaks fromcausing spurious transitions in the output signal at D_(OUT).

FIG. 5 also shows greater detail of an example of an output edgetriggered disable one-shot. One input terminal of exclusive-OR gate 852receives the output signal directly while the other input terminal iscoupled to the output signal through resistor 756 and to ground throughcapacitor 854. Resistor 856 and capacitor 854 further delay the outputsignal such that, when a pulse edge occurs in the output signal, the twoinputs of XOR 852 will be at different values for a time perioddetermined by the RC constant of resistor 856 and capacitor 854, whichtherefore also determine the width of the disable pulses generated byXOR 852.

Another embodiment of a receiver circuit suitable for use as receiver220 of FIG. 2 is receiver circuit 920 of FIG. 6. The high amplitude ofthe feedback pulses contributes to ringing, overshoot and undershoot atthe output of bandpass filter 834 of receiver 820. Thus, receiver 920 isconstructed with signal disable switch 948 interposed between the outputof input amplifier 926 and bandpass filter 934. This configurationpermits the disable pulses generated by XOR 952 responsive to the edgesin the output signal to isolate bandpass filter 934, comparator 942 andAGC peak detector 936 from the receive signal path when feedback fromD_(OUT). This configuration causes the feedback transient to settle morerapidly because the transient is prevented from entering the bandpassfilter 934 where the transient is prolonged due to the increase inringing, overshoot and undershoot that would otherwise occur due tofeedback transients in the output response of bandpass filter 934. Thispermits receiver 920 to tolerate larger amplitude feedback signals, usea narrower bandwidth to improve the signal to noise ratio of thereceiver, or operate at faster pulse rates without corruption of thedata output signal at D_(OUT).

Because of variations in device tolerances and operating conditions, itis not possible to exactly predict when the feedback transients in thereceived signal path will settle. For slower communications formats, theduration of the disable pulses may be extended to account for variationsin the settling time of the feedback peaks. However, fastercommunications formats, such as the 4 MB format described above, havenarrow windows because the data is related to the temporal position ofthe pulse which requires rapid settling times. As a result, the disablesignal generated by XOR 952 may not coincide exactly with the feedbackpeaks.

For example, if the disable signal is 1.5 us, then valid input signaltransitions which have less than 1.5 us between them cannot be detected.To capture these signal transitions, it is necessary to set the disablesignal duration to the minimum required to prevent feedback disruption.However, due to variances in IC timing circuit tolerances and variancesin feedback due to variances in receiver packages and circuit boardtrace layout, it becomes necessary to set the signal disable period to alarger value than is typically required so as to ensure that mostreceivers will function without feedback disruption. This adds adifficult engineering burden of correctly determining the optimum signaldisable duration and undesirable limiting maximum pulse rate on receiverpackages or board layouts which have low feedback levels.

Yet another embodiment of a receiver circuit suitable for use asintegrated receiver 220 of FIG. 2 is receiver 1020 of FIG. 7. In orderto accommodate faster communications formats and address variation insignal settling, receiver 1020 is designed with positive feedback andfeedback detection which monitors the feedback transients and preventssignal disable switch 1048 from closing while a feedback transient ispresent.

Feedback detect comparator 1060 compares the amplified input data signalat the output of input amplifier 1026 to the detection threshold voltagelevel V_(DET) and output a high level signal so long as the amplifiedinput signal is greater that the detection threshold. Exclusive-OR 1062compares the output of the feedback detect comparator 1060 with thedelayed digital output signal and outputs a high level signal if the twosignals differ. Conversely, exclusive-OR 1062 outputs a low level if thetwo signals are in agreement. This low output signal will propagatethrough AND gate 1064 to close signal disable switch 1048 before theedge triggered disable signal output from exclusive-OR 1052 wouldnormally cause switch 1048 to close. This permits receiver 1020 tooperate at higher speeds when the transient settling time is faster thanthat predicted solely by the timing of the edge-triggered one shotcircuit.

The photodiode of the present invention can also be incorporated intodifferential methods for receiving an infrared signal. In FIG. 8, a pairof photodiodes 1130 and 1140 are separately input to differential inputsD_(IR1) and D_(IR2), respectively, of receiver 1120. Photodiode 1130 isformed by producing N-diffusion 1132 in P-substrate 210. Similarly,photodiode 1140 is formed by producing N-diffusion 1142 in substrate1110. But photodiode 1140 is also covered with an opaque shield orcoating 1146 which prevents the infrared light signal from reachingphotodiode 1140. Photodiodes 1130 and 1140 are fabricated using highprecision semiconductor fabrication techniques and therefore can beproduced with a high level of symmetry with respect to the receiveroutput terminal D_(OUT) so as to have almost identical feedbackcapacitance to D_(OUT). As a result, excellent immunity to externalnoise sources and receiver feedback can be obtained using the matchedphotodiodes 1130 and 1140.

The receiver circuits 420, 620, 820, 920 and 1020 can each be modifiedto operate in a differential mode with the photodiodes illustrated inFIG. 8, as would be apparent to one skilled in the art.

Having illustrated and described the principles of the present inventionin the context of the embodiments described above, it should be readilyapparent to those skilled in the art that the invention can be modifiedin arrangement and detail without departing from such principles. Forexample, while much of the circuitry described herein is constructedusing analog circuits, such as the analog timer and comparators, itshould be readily understood that similar function can be obtained usingdigital components.

I claim:
 1. An integrated photodiode and receiver circuit on asubstrate, the integrated receiver circuit comprising: a first diffusedregion in the substrate for receiving an input signal; a first circuitinput terminal coupled to the first diffused region; a circuit outputterminal; an input amplifier interposed between the first circuit inputand receiver output terminals, the input amplifier being configured toreceive and amplify the input signal to produce an amplified inputsignal, and wherein the input amplifier is further configured to varythe gain of the input amplifier responsive to a gain control signal; abandpass filter interposed between the input amplifier and the circuitoutput terminal, the bandpass filter being configured to receive andbandpass filter the amplified input signal so as to produce a filteredinput signal; a comparator interposed between the bandpass filter andthe circuit output terminal, the comparator being configured to comparethe filtered input signal to a detection threshold voltage level inorder to generate a digital output signal; a delay circuit interposedbetween the comparator and the circuit output terminal, the delaycircuit being configured to receive the digital output signal andgenerate a delayed digital output signal responsive thereto; anautomatic gain control circuit configured to receive the filtered inputsignal, the automatic gain control circuit comparing the filtered inputsignal to an automatic gain control threshold voltage and generating thegain control signal responsive thereto; an isolation control signalgenerator configured to receive the delayed digital output signal andgenerating an isolation control signal responsive thereto; and anisolation switch interposed between the input amplifier and theautomatic gain control circuit, the isolation switch being configured toreceive the isolation control signal and, responsive thereto, isolatethe automatic gain control circuit from the amplified input signal. 2.The integrated receiver circuit of claim 1 wherein the isolation switchis further configured to isolate the comparator from the amplified inputsignal responsive to the isolation control signal.
 3. The integratedreceiver circuit of claim 1 wherein the isolation switch is furtherconfigured to isolate the bandpass filter from the amplified inputsignal responsive to the isolation control signal.
 4. The integratedreceiver of claim 1 wherein the isolation control signal generatorcomprises a one-shot circuit configured to generate an output pulse of apredetermined duration responsive to a falling edge in the delayeddigital output signal.
 5. The integrated receiver of claim 1 wherein theisolation control signal generator comprises an edge triggered one-shotcircuit configured to generate an output pulse of a predeterminedduration responsive to each edge in the delayed digital output signal.6. The integrated receiver of claim 5 wherein the edge triggeredone-shot circuit comprises: a first exclusive-OR gate having first andsecond input terminals and an output terminal, wherein the first inputterminal is configured to receive the delayed digital output signal; aresistor having first and second terminals, wherein the first terminalis configured to receive the delayed digital output signal, and furtherwherein the second terminal is coupled to the second input terminal ofthe first exclusive-OR gate; and a capacitor coupled between the secondinput terminal of the first exclusive-OR gate and ground; whereby thefirst exclusive-OR gate generates the isolation control signal at itsoutput terminal responsive to the delayed digital output signal.
 7. Theintegrated receiver of claim 5 wherein the isolation control signalgenerator further includes: a feedback detect comparator configured toreceive the amplified input signal and compare it to the detectionthreshold voltage level to produce a feedback detect signal; a secondexclusive-OR gate having first and second input terminals and an outputterminal, wherein the first input terminal is configured to receive thefeedback detect signal and the second input terminal is configured toreceive the delayed digital output signal; and an AND gate having firstand second input terminals and an output terminal, wherein the firstinput terminal is coupled to the output terminal of the firstexclusive-OR gate and the second input terminal is coupled to the outputterminal of the second exclusive-OR gate whereby the AND gate generatesthe isolation control signal at its output terminal.
 8. An infraredreceiver circuit formed on a substrate, the circuit comprising: a firstdiffusion in the substrate; an input amplifier having first and secondinput terminals, a gain control terminal and an output terminal, thefirst input terminal being configured to receive a first bias voltageand the second input terminal being coupled to the first diffusion; abandpass filter having input and output terminals, the input terminalbeing coupled to the output terminal of the input amplifier; acomparator having first and second input terminals and an outputterminal, the first input terminal being configured to receive adetection threshold voltage, the second input terminal being coupled tothe output of the bandpass filter, and the output terminal being coupledto an output terminal of the receiver circuit; an automatic gain controlcircuit having first and second input terminals and an output terminal,the first input terminal being coupled to the output of the bandpassfilter, the second input terminal being configured to receive anautomatic gain control threshold voltage, and the output terminal beingcoupled to the gain control terminal of the input amplifier; a delaycircuit having input and output terminals, wherein the input terminal iscoupled to the output terminal of the comparator such that a delayeddigital output signal is generated at the output terminal of the delaycircuit responsive to an input data signal received at the inputterminal of the delay circuit; an isolation control signal generatorhaving input and output terminals, wherein the input terminal is coupledto the output terminal of the delay circuit, and wherein the isolationcontrol signal generator is configured to generate a pulse ofpredetermined duration at its output terminal responsive to an edge inthe delayed digital output signal; and an isolation switch having input,output and control terminals, the control terminal being coupled to theoutput terminal of the isolation control signal generator such that theisolation switch isolates the input terminal thereof from the outputterminal thereof responsive to the automatic gain control signal, theinput terminal of the isolation switch being coupled to the outputterminal of the input amplifier, the output terminal of the isolationswitch being coupled to the first input terminal of the automatic gaincontrol circuit such that the isolation switch is interposed between thefirst input terminal of the automatic gain control circuit and theoutput terminal of the input amplifier whereby the isolation switchisolates the automatic gain control circuit from the output terminal ofthe input amplifier responsive to the isolation control signal.
 9. Thereceiver of claim 8 wherein the output terminal of the isolation switchis further coupled to the second input terminal of the comparator suchthat the isolation switch is interposed between the second inputterminal of the comparator and the output terminal of the inputamplifier whereby the isolation switch isolates the comparator from theoutput terminal of the input amplifier responsive to the isolationcontrol signal.
 10. The receiver of claim 8 wherein the output terminalof the isolation switch is further coupled to the input terminal of thebandpass filter such that the isolation switch is interposed between theinput terminal of the bandpass filter and the output terminal of theinput amplifier whereby the isolation switch isolates the bandpassfilter from the output terminal of the input amplifier responsive to theisolation control signal.
 11. The receiver of claim 9 wherein the outputterminal of the isolation switch is further coupled to the inputterminal of the bandpass filter such that the isolation switch isinterposed between the input terminal of the bandpass filter and theoutput terminal of the input amplifier whereby the isolation switchisolates the bandpass filter from the output terminal of the inputamplifier responsive to the isolation control signal.
 12. The receiverof claim 11 wherein the isolation control signal generator comprises aone-shot circuit configured to generate an output pulse of apredetermined duration at the output terminal responsive to a fallingedge in the delayed digital output signal.
 13. The receiver of claim 11wherein the isolation control signal generator comprises an edgetriggered one-shot circuit configured to generate an output pulse of apredetermined duration at the output terminal responsive to each edge inthe delayed digital output signal.
 14. The receiver of claim 13 whereinthe edge triggered one-shot circuit comprises: a first exclusive-OR gatehaving first and second input terminals and an output terminal, thefirst input terminal being coupled to the output terminal of the delaycircuit and the output terminal being coupled to the control terminal ofthe isolation switch; a resistor having first and second terminals, thefirst terminal being coupled to the output terminal of the delaycircuit, and the second terminal being coupled to the second inputterminal of the first exclusive-OR gate; and a capacitor coupled betweenthe second input terminal of the first exclusive-OR gate and ground;whereby the first exclusive-OR gate generates the isolation controlsignal at its output terminal responsive to the delayed digital outputsignal.
 15. The receiver of claim 14 wherein the isolation controlsignal generator further includes: a feedback detect comparator havingfirst and second input terminals and an output terminal, the first inputterminal being coupled to the output terminal of the input amplifier,and the second input terminal being configured to receive the detectionthreshold voltage; a second exclusive-OR gate having first and secondinput terminals and an output terminal, the first input terminal beingcoupled to the output terminal of the feedback detect comparator, andthe second input terminal being coupled to the output terminal of thedelay circuit; and an AND gate having first and second input terminalsand an output terminal, the first input terminal being coupled to theoutput terminal of the first exclusive-OR gate, the second inputterminal being coupled to the output terminal of the second exclusive-ORgate, and the output terminal being coupled to the control terminal ofthe isolation switch.
 16. The infrared receiver circuit of claim 8, thecircuit further comprising: a second diffusion in the substrate; anopaque shield formed over the second diffusion and spaced apart from thefirst diffusion; and wherein the second diffusion is coupled to thefirst input terminal of the input amplifier.
 17. A method forsuppressing feedback in a photodiode and a receiver fabricated on asubstrate, the method comprising the steps: receiving an input signalwith the photodiode; amplifying the input signal to produce an amplifiedinput signal; controlling the gain of the input signal amplificationresponsive to the magnitude of the amplified input signal; comparing theamplified input signal to a detection threshold voltage to produce adigital output signal; and holding the gain at a substantially constantlevel in response to a fast signal transition in the digital outputsignal.
 18. The method of claim 17 including the step of filtering theamplified input signal to produce a filtered input signal.
 19. Themethod of claim 17 wherein: the step of controlling the gain of theinput signal amplification responsive to the magnitude of the amplifiedinput signal includes the step of controlling the gain using anautomatic gain control circuit; and the step of holding the gain at asubstantially constant level includes the steps: generating an isolationsignal responsive to the digital output signal, and isolating theautomatic gain control circuit from the amplified input signalresponsive to the isolation signal.
 20. The method of claim 19 whereinthe step of comparing the amplified input signal to a detectionthreshold voltage includes using a comparator circuit to compare theamplified input signal to the detection threshold voltage and furtherincluding the step of isolating the comparator from the amplified inputsignal responsive to the isolation signal.
 21. The method of claim 19including: filtering the amplified input signal using a bandpass filterto produce a filtered input signal; and isolating the bandpass filterfrom the amplified input data signal responsive to the isolation signal.22. The method of claim 19 wherein the step of generating an isolationsignal includes generating a pulse of a predetermined durationresponsive to each falling edge in the digital output signal.
 23. Themethod of claim 19 wherein the step of generating an isolation signalincludes generating a pulse of a predetermined duration responsive toeach edge in the digital output signal.
 24. The method of claim 17wherein the step of comparing the filtered input signal to a detectionthreshold voltage to produce a digital output signal includes the stepof introducing a time delay to the digital output signal.